Calibrating RF path delay and IQ phase imbalance for polar transmit system

ABSTRACT

A method of calibrating parameters for a polar transmitter (Polar TX) system includes receiving phase information derived from transmission information in a Polar TX for producing a radio frequency (RF) broadcast signal. An Inphase local oscillator (LO_I) signal and a quadrature phase local oscillator (LO_Q) signal are derived from a combination of a first signal and the phase information using a digital phase lock loop. A feedback receiver (FBR) receives the RF broadcast signal provided by the Polar TX. The LO_I signal and the LO_Q signal are mixed with the RF broadcast signal to obtain mixer output signals. RF path delay and IQ phase imbalance are concurrently determined as a function of the first signal and of the mixer output signals.

TECHNICAL FIELD

The present subject matter generally relates to communicationarchitectures and, in particular, to apparatus and methods for measuringradio frequency (RF) path signal delay and Inphase-Quadrature phase (IQ)phase imbalance in a polar transmitter (Polar TX) system.

BACKGROUND

Polar Transmitter (Polar TX) architectures are very attractive formodern radios because such architectures can provide improved area andpower consumption characteristics compared with conventional analogarchitectures. One drawback to a polar system containing a Polar TX anda feedback receiver (FBR) system is that the phase modulation is notcancelled perfectly inside the FBR due to the inherent delay in the RFpath, sometimes called RF delay path delay. When RF path delay is known,phase cancelation can be achieved by digital post processing. It isdesirable to calibrate the RF path delay, and also be able to calibratethe IQ phase imbalance, in order to facilitate successful postprocessing.

BRIEF DESCRIPTION OF THE DRAWINGS

In the drawings, which are not necessarily drawn to scale, like numeralsmay describe similar components in different views. Like numerals havingdifferent letter suffixes may represent different instances of similarcomponents. The drawings illustrate generally, by way of example, butnot by way of limitation, various embodiments discussed in the presentdocument.

FIG. 1 illustrates an overview of a system containing a Polar TX and anFBR, according to an embodiment.

FIG. 2 illustrates an IQ constellation with a single point, according toan embodiment.

FIG. 3 is an illustration of the effect of IQ phase imbalance on the IQconstellation diagram of FIG. 2, according to an embodiment.

FIG. 4A is a representation of the constellation diagram of FIG. 2 bothwith and without phase error, according to an embodiment

FIG. 4B illustrates the deviation of the amplitude squared from theaverage amplitude squared, in terms of receiver power, according to anembodiment

FIG. 4C illustrates a phase imbalance detector characteristic, accordingto an embodiment.

FIG. 4D illustrates the accuracy of detected phase imbalance, accordingto an embodiment.

FIG. 5 is a flow chart illustrating a method using various measurementsto provide RF path signal delay and IQ phase imbalance, according to anembodiment.

FIG. 6 is a functional diagram of a portion of an end-to-end networkarchitecture of an LTE network in accordance with some embodiments.

FIG. 7 illustrates a functional block diagram of user equipment (UE) inaccordance with some embodiments.

DETAILED DESCRIPTION

An FBR with a modulated local oscillator (LO) should cancel the phasemodulation and deliver only the amplitude demodulation of the RF signal.However, as a first problem, experience with Polar TXs and FBRs hasshown that the phase modulation is not cancelled perfectly inside theFBR due to inherent RF path delay. When the RF path delay is known, thephase cancellation can be achieved by digital post-processing. But inorder to accomplish this it is important to solve the first problem byknowing how to calibrate the RF path delay.

A second problem, the FBR has limited accuracy due to the IQ phaseimbalance (i.e., non-perfect ninety degrees between the local oscillatorfor the Inphase (LO_I) and the local oscillator for the quadrature phase(LO_Q) signals. It is important to resolve the second problem bycalibrating the IQ phase imbalance and, once the IQ phase imbalance iscalibrated or known, compensating for it in the digital signalprocessing of the FBR.

Phase imbalance calibration is well known for a standard IQ receiver,where the LO signal is unmodulated. An external RF signal with aconstant envelope modulation such as(RF_sig=cos(j*phi_RF+j*2*pi*Fsin*t)) can be injected into the FBR input.At the same time the phase modulation of the internal LO signal can bedisabled, and the transmitter output power can be disabled by switchingoff the power amplifier. By doing this the receiver behaves like astandard IQ receiver with an unmodulated LO. A disadvantage is that thismethod requires an external signal generator and is slower than desiredsince it involves control of on-chip components, and also control ofchip-external components such as a signal generator. The disclosedmethod does not require an external device to be controlled. Thecomplete procedure can be integrated on-chip.

Further, such RF path delay calibration is done in the lab by extensivemeasurements, storing the results in a non-volatile memory as acalibration data set. In this case, any deviation of RF delay from partto part in the production had to be budgeted.

Another approach for the calibration of the RF path delay requiredeither dedicated hardware or complex calculations, storing the receivedsamples inside RAM and perform cross-correlation or least-mean-squaresearch and other techniques or hardware.

While the IQ phase imbalance requirement can be calibrated in anIQ-based architecture, as above, it has not been possible to apply sucha calculation for polar architecture. Also until now, the IQ phaseimbalance had to be achieved by a circuit design. This has drawbacks incurrent consumption and complexity. Further, the maximum frequency ofthe LO path was limited for previous calibration methods. The disclosedembodiments enable obtaining the RF path delay and the IQ phaseimbalance at essentially the same time, thus saving calibration time.The RF path delay is a side-product of calibrating the IQ phaseimbalance. Moreover, the calculation method for the embodiments is verysimple, does not require additional hardware beyond the transmitter andthe receiver, and can be applied even if no IQ phase imbalancecalibration is required. The disclosed embodiments allow an increased IQphase imbalance inside a LO path for FBR for a polar architecture,Further, the limitation of maximum frequency of the LO path for previouscalibration methods can be relaxed by enabling calibration of IQ phaseimbalance in a polar system as disclosed herein. Some embodiments areapplicable to polar architectures, although the scope of the embodimentsis not limited in this respect, as some embodiments may be applicable toother architectures including, but not limited to IQ architectures.

FIG. 1 illustrates an overview of a system comprising a Polar TX and anFBR. The Polar TX 100 can receive transmission information such as TXsignal 101 in the form of polar symbols or polar information includingamplitude information (A) and phase information (φ). In some examples, abaseband modulator 103 can provide the transmission information inCartesian symbols (I, Q). The Polar TX 100 can include a coordinaterotation digital computer (CORDIC) 105 to translate the input Cartesiansymbols (I, Q) to polar symbols, or information, designated as amplitudemodulation and phase modulation (AM, PM) at 107, 109, respectively.Phase modulation PM may be referred to herein as phase information (φ).

In certain examples, the Polar TX 100 includes a phase processing paththat includes digital phase lock loop (DPLL) 113 to provide the phaseinformation of the carrier signal based on the phase information PM, orfrequency information derived from the phase information PM viadifferentiator 111. In some embodiments, combiner 112 is included toallow a signal, FM_tune 110, discussed subsequently, to be injected inthe system for use in RF path delay calibration and phase imbalancecalculation.

-   -   The Polar TX 100 may also include an amplitude processing path        beginning at 107 for amplitude information. AM information at        107 is converted to analog via digital to analog convertor (DAC)        108. In some examples, the phase processing path and the        amplitude processing path can each include an interpolator        circuit, not shown, for up-sampling or down-sampling the phase,        frequency or amplitude information. In certain examples, the        phase modulated carrier signal output from the DPLL 113 and the        processed amplitude information from DAC 108 can be combined        using the combiner 115 to provide an RF transmission signal for        broadcast using the antenna 123. The radio transmission signal        may be amplified in power amplifier 117. Duplexer 119 and        coupler 121 may couple the RF transmission signal to the antenna        123. Duplexer (or any kind of filter) introduces a signal        propagation delay, referred to as Δt in FIG. 1, Δt symbolizing        the RF path delay.

The FBR 102 includes line 120 for receiving RF signal PM (t+Δt) on line120, which is coupled to mixers 122, 124. The LO_I signal from DPLL 113is coupled to mixer 122 via line 118 while the LO_Q signal is coupledover line 119 to mixer 124. Item 116 symbolizes the fact that, asmentioned above, there is IQ phase imbalance, namely a non-perfectninety degrees between the LO_I and the LO_Q signals, this IQ phaseimbalance being indicated by the symbol “e” at 116, to be discussedsubsequently. One possible and often used method to generate signalsLO_Q and LO_I from the LO output of DPPL 113 is to use a frequencydivider, not shown. The LO signal output of the DPLL 113 may be dividedby 2 or 4. The output of this divider can generate LO_I at line 118 anda 90°-shifted (with error e) LO_Q signal at 119. Other methods may beused to achieve this signal separation of LO_I and LO_Q from the LOoutput signal from DPPL 113. The output of mixer 122 is the RF signalfrom line 120 (PM (t+Δt)) combined with the LO_I signal, and the outputof mixer 124, is the RF signal from line 120 (PM(t+Δt)) combined withthe LO_Q signal. Mixers 122 and 124 receive signals LO_I and LO_Q fromDPLL 113 in order to frequency-shift the tapped transmit signal fromline 120 to baseband frequency. The outputs of mixers 122 and 124 arerespectively coupled to an ADC for Inphase information (ADC-I) 128 andan ADC for quadrature information (ADC-Q) 129. Each of ADC-I 128 andADC-Q 129 is coupled to FBR processing circuitry 130. FBR processingcircuitry 130 may include a CORDIC (not shown) that operates much likeCORDIC 105 in that it converts the IQ representation of the tapped anddown-mixed transmit signal to a polar representation having an amplitudecomponent AM and a phase component PM, each of which are respectivelyshown as AM information signal AM_FBR 132 and phase information signalPM (t+Δt)-PM(t) 134. These signals may be used for FBR functions such asthose listed at 136 for desired use. AM information AM_FBR 132 and phaseinformation, PM (t+Δt)-PM(t) 134, are also used to calculate RF pathdelay and to calibrate IQ phase imbalance as discussed below.

The disclosed implementation can be explained in terms of a well-knownI/Q constellation diagram. The symbols on a constellation diagram may berepresented as complex numbers and can be visualized as points on thecomplex plane. The real and imaginary axes are often called the Inphase, or I-axis, and the Quadrature phase, or Q-axis, respectively.Plotting several symbols in a scatter diagram produces a constellationdiagram with a number of points around the circle of the diagram. FIG. 2illustrates an IQ constellation with a single point, according to anembodiment. This illustration is a specific example of a generalconstellation diagram and is helpful in understanding how the disclosedembodiments vary. Thus, any point in the constellation diagram of FIG. 2will fall on a simple straight line on an IQ-diagram (to be discussedsubsequently) if there is no RF path delay Δt. The constellation pointsof a general constellation diagram will, at the FBR input, be reduced toa single point as shown in FIG. 2, if the amplitude modulation isdisabled (i.e., the points of the constellation diagram are just goingin a circle). The position of the resulting point of FIG. 2 is thereforeseen to be dependent on the RF path delay.

RF Path Delay

When an FM signal such as FM_tune 110 in FIG. 1 is inserted into thetransmitter via combiner 112 in an embodiment, the output of DPPL 113will include FM_tune as part of LO_Q and LO_I over lines 119 and 118,respectively.

The LO_I and LO_Q signals at 118 and 119, respectively, are combined viacombiners 122 and 124, respectively, with the received RF input signalfrom antenna 123 on line 120. The output of the FBR may therefore beviewed as an IQ constellation with a single point. The position of thispoint depends on the RF path delay, which is to be calculated, and canbe shifted around the circle 200 of FIG. 2, such as at 201, 203, if theRF delay is changed or if the modulation frequency is changed. Thepoints on the circle 200 of FIG. 2 may be given by the followingequations:M_point_1=(2*n*f _(RF) *Δt)  (1)When no FM_tune is applied (i.e., at 201).M_point_2=(2*n*f _(RF) *Δt)+(2*n*FM_tune*Δt)  (2)When FM_tune is applied (i.e., at 203).delta_M=M_point_2−M_point_1  (3)Out of these equations Δt can be calculated as:Δt=delta_M/(2*n*FM_tune)  (4)

Where Δt is the RF path delay

IQ Phase Imbalance

FIG. 4A is a representation of the constellation diagram of FIG. 2 bothwith and without phase error, according to an embodiment. Theconstellation diagram without IQ phase error is seen at 410 of FIG. 4A,while the effects of phase error is seen at 420, where the I/Q phase isnot precisely ninety degrees. FIG. 4B illustrates the deviation of theamplitude signal squared from the average amplitude signal squared, interms of receiver power, according to an embodiment. Power is discussedwith respect to Equations (5)-(9) below) for points on curve 420 of FIG.4A.

FIG. 3 is an illustration of the effect of IQ phase imbalance on the IQconstellation diagram of FIG. 2, according to an embodiment. The IQphase imbalance makes an ellipse out of the IQ constellation circle 200of FIG. 2. This ellipse effect is seen in FIG. 3 where circle 300becomes ellipse ellipse 300 ¹. When the FM_tune signal injected at 110is increased, and the amplitude A of signal 310 may be monitored, whichmay be at point 132 in FIG. 1. The amplitude is seen to become asinus-like trajectory 310 about the average amplitude A_(aver) in FIG.3, where points 301 ¹, 303 ¹, 305 ¹ and 307 ¹ correspond to points 301,303, 305, and 307. A 45° rotation of the measured points can be achievedby proper selection of FM_tune. Increasing the FM_tune will, in thedescribed embodiment, lead to a rotation of the measured pointcounterclock-wise, decreasing the FM_tune will lead to a clock-wiseshift of the measured point. The FM_tune may be tuned and the phase ofthe new point may be compared to the last measured point. This should berepeated until 45 degree shift between the points is achieved. Thisprocedure is illustrated in the loop at 520, 530, 540 and 550 in FIG. 5,discussed below. This would give, in FIG. 3, a whole period of theamplitude signal after 4 measurement points A_(n) with n=1 . . . 4,symbolized at points 301, 303, 305, and 307, respectively. The IQ phaseimbalance |e|, seen symbolically as a divergence from ninety degrees at116 of FIG. 1, can be calculated out of these 4 measured amplitudepoints. The sign of the phase imbalance is given by the comparison ofthe A_(n) measured at quadrant I or III of FIG. 3. The sign may be named(A_(n) _(_)equal_IQsign) with the A_(n) measured at quadrant II or IV ofFIG. 3 being named (=A_(n) _(_)nonequal_IQsign). If An_equal_IQsign isgreater than An_nonequal_IQsign, then the phase error e was a positivenumber, otherwise negative. The formula for calculating the magnitude|e| is:

$\begin{matrix}{P_{aver} = {\frac{1}{4}{\sum\limits_{n = 1}^{4}A_{n}^{2}}}} & (5) \\{P_{{norm}\; 1} = {A_{n = 1}^{2} - P_{aver}}} & (6) \\{P_{{norm}\; 2} = {A_{n = 2}^{2} - P_{aver}}} & (7) \\{P_{\max} = \sqrt[2]{P_{{norm}\; 1}^{2} + P_{{norm}\; 2}^{2}}} & (8) \\{{{\mathbb{e}}} \approx \frac{P_{\max}}{P_{aver}}} & (9)\end{matrix}$

The phase calculation is not precise for bigger phase errors. In orderto simplify the mathematics the Acos and Asin functions were avoided,but the accuracy remains better than 0.5°, which is sufficient for phaseimbalances smaller than 20°.

FIG. 4C illustrates a phase imbalance detector characteristic, accordingto an embodiment, while FIG. 4D illustrates the accuracy of the detectedphase imbalance, according to an embodiment. The accuracy may be seenfrom FIGS. 4C and 4D. Phase imbalance has been measured to determineaccuracy of the above method. As illustrated in FIG. 4C, the measuredphase imbalance in degrees is very close to the applied phase imbalancein degrees. Laboratory measurements have shown the error to be within0.5°. FIG. 4D illustrates measurement error as a function of appliedphase imbalance

FIG. 5 is a flow chart illustrating a method using various measurementsto provide the RF path delay and the IQ phase imbalance, according to anembodiment. At 510 the signal FM_tune at 110 of FIG. 1 is set to zeroand measurements of phase modulation and amplitude modulation are takenat 132, 134, respectively, of FIG. 1. This is taken at n=1 (301 ¹ ofFIG. 3) and N is then increased by I such that n=2 and FM_tune isincreased at 520. At 530 the phase modulation information is measured asat 134 of FIG. 1. This is the measurement n=2 (303 ¹ of FIG. 3). At 540a test is taken to determine whether the difference in phasemeasurements is less than forty-five degrees. If the answer is NO, thenthe FM_tune signal is increased at 520 and the process continues. If theanswer is YES, then a test is made to determine whether N is equal to orgreater than 4. If the answer is NO, then N is increased by 1 and theprocess continues at 520. At the same time, the RF delay can becalculated at 560 using Equations (1) through (4), above. Thiscalculation need be taken only the first time the NO answer to test 550is taken, and the RF delay is valid regardless of the number of timesthe loop 520, 530, 540, 550 is performed.

The loop 520, 530, 540, 550 is continued until the answer at test 550 isYES. When the YES decision is taken, at 570 the magnitude |e| of the IQphase imbalance may be calculated using Equations (5) through (9). Thephase error |e| will be positive if An_equal_IQsign>An_nonequal_IQsign,otherwise negative, as discussed above. The determined RF path delay andIQ phase imbalance at 560 and 570, respectively, may be used forcorrection purposes as discussed above. Although the parameters may bedetermined independently and/or concurrently, this is not a requirementand the scope of the embodiments is not limited in this respect.

FIG. 6 shows a portion of an end-to-end network architecture of an LTEnetwork with various components of the network in accordance with someembodiments. The network 600 comprises a radio access network (RAN)(e.g., as depicted, the E-UTRAN or evolved universal terrestrial radioaccess network) 601 and the core network 620 (e.g., shown as an evolvedpacket core (EPC)) coupled together through an S1 interface 615. Forconvenience and brevity sake, only a portion of the core network 620, aswell as the RAN 600, is shown.

The core network 620 includes mobility management entity MME servinggateway (serving GW) 624, and packet data network gateway (PDN GW) 626.The RAN includes enhanced node B's (eNBs) 604 (which may operate as basestations) for communicating with user equipment (UE) 602. The eNBs 604may include macro eNBs and low power (LP) eNBs.

The MME is similar in function to the control plane of legacy ServingGPRS Support Nodes (SGSN). The MME manages mobility aspects in accesssuch as gateway selection and tracking area list management. The servingGW 624 terminates the interface toward the RAN 600, and routes datapackets between the RAN 600 and the core network 620. In addition, itmay be a local mobility anchor point for inter-eNB handovers and alsomay provide an anchor for inter-3GPP mobility. Other responsibilitiesmay include lawful intercept, charging, and some policy enforcement. Theserving GW 624 and the MME may be implemented in one physical node orseparate physical nodes. The PDN GW 626 terminates an SGi interfacetoward the packet data network (PDN). The PDN GW 626 routes data packetsbetween the EPC 620 and the external PDN, and may be a key node forpolicy enforcement and charging data collection. It may also provide ananchor point for mobility with non-LTE accesses. The external PDN can beany kind of IP network, as well as an IP Multimedia Subsystem (IMS)domain. The PDN GW 626 and the serving GW 624 may be implemented in onephysical node or separated physical nodes.

The eNBs 604 (macro and micro) terminate the air interface protocol andmay be the first point of contact for a UE 602. In some embodiments, aneNB 604 may fulfill various logical functions for the RAN 600 includingbut not limited to RNC (radio network controller functions) such asradio bearer management, uplink and downlink dynamic radio resourcemanagement and data packet scheduling, and mobility management. Inaccordance with embodiments, UEs 602 may be configured to communicateOFDM communication signals with an eNB 604 over a multicarriercommunication channel in accordance with an OFDMA communicationtechnique. The OFDM signals may comprise a plurality of orthogonalsubcarriers.

The S1 interface 615 is the interface that separates the RAN 600 and theEPC 620. It is split into two parts: the S1-U, which carries trafficdata between the eNBs 604 and the serving GW 624, and the S1-MME, whichis a signaling interface between the eNBs 604 and the MME. The X2interface is the interface between eNBs 604. The X2 interface comprisestwo parts, the X2-C and X2-U. The X2-C is the control plane interfacebetween the eNBs 604, while the X2-U is the user plane interface betweenthe eNBs 604.

With cellular networks, LP cells are typically used to extend coverageto indoor areas where outdoor signals do not reach well, or to addnetwork capacity in areas with very dense phone usage, such as trainstations. As used herein, the term low power (LP) eNB refers to anysuitable relatively low power eNB for implementing a narrower cell(narrower than a macro cell) such as a femtocell, a picocell, or a microcell. Femtocell eNBs are typically provided by a mobile network operatorto its residential or enterprise customers. A femtocell is typically thesize of a residential gateway or smaller, and generally connects to theuser's broadband line. Once plugged in, the femtocell connects to themobile operator's mobile network and provides extra coverage in a rangeof typically 30 to 50 meters for residential femtocells. Thus, a LP eNBmight be a femtocell eNB since it is coupled through the PDN GW 626.Similarly, a picocell is a wireless communication system typicallycovering a small area, such as in-building (offices, shopping malls,train stations, etc.), or more recently in-aircraft. A picocell eNB cangenerally connect through the X2 link to another eNB such as a macro eNBthrough its base station controller (BSC) functionality. Thus, LP eNBmay be implemented with a picocell eNB since it is coupled to a macroeNB via an X2 interface. Picocell eNBs or other LP eNBs may incorporatesome or all functionality of a macro eNB. In some cases, this may bereferred to as an access point base station or enterprise femtocell.

In some embodiments, a downlink resource grid may be used for downlinktransmissions from an eNB to a UE. The grid may be a time-frequencygrid, called a resource grid, which is the physical resource in thedownlink in each slot. Such a time-frequency plane representation is acommon practice for OFDM systems, which makes it intuitive for radioresource allocation. Each column and each row of the resource gridcorrespond to one OFDM symbol and one OFDM subcarrier, respectively. Theduration of the resource grid in the time domain corresponds to one slotin a radio frame. The smallest time-frequency unit in a resource grid isdenoted as a resource element. Each resource grid comprises a number ofresource blocks, which describe the mapping of certain physical channelsto resource elements. Each resource block comprises a collection ofresource elements and in the frequency domain, this represents thesmallest quanta of resources that currently can be allocated. There areseveral different physical downlink channels that are conveyed usingsuch resource blocks. With particular relevance to this disclosure, twoof these physical downlink channels are the physical downlink sharedchannel and the physical down link control channel.

The physical downlink shared channel (PDSCH) carries user data andhigher-layer signaling to a UE 602 (FIG. 6). The physical downlinkcontrol channel (PDCCH) carries information about the transport formatand resource allocations related to the PDSCH channel, among otherthings. It also informs the UE about the transport format, resourceallocation, and H-ARQ information related to the uplink shared channel.Typically, downlink scheduling (assigning control and shared channelresource blocks to UEs within a cell) is performed at the eNB based onchannel quality information fed back from the UEs to the eNB, and thenthe downlink resource assignment information is sent to a UE on thecontrol channel (PDCCH) used for (assigned to) the UE.

The PDCCH uses CCEs (control channel elements) to convey the controlinformation. Before being mapped to resource elements, the PDCCHcomplex-valued symbols are first organized into quadruplets, which arethen permuted using a sub-block inter-leaver for rate matching. EachPDCCH is transmitted using one or more of these control channel elements(CCEs), where each CCE corresponds to nine sets of four physicalresource elements known as resource element groups (REGs). Four QPSKsymbols are mapped to each REG. The PDCCH can be transmitted using oneor more CCEs, depending on the size of DCI and the channel condition.There may be four or more different PDCCH formats defined in LTE withdifferent numbers of CCEs (e.g., aggregation level, L,=1, 2, 4, or 8).

FIG. 7 illustrates a functional block diagram of a UE in accordance withsome embodiments. The UE 700 may be suitable for use as any one or moreof the UEs 602 illustrated in FIG. 6. The UE 700 may include physicallayer circuitry 702 for transmitting and receiving signals to and fromeNBs 604 (FIG. 6) using one or more antennas 701. UE 700 may alsoinclude medium access control layer (MAC) circuitry 704 for controllingaccess to the wireless medium. UE 700 may also include processingcircuitry 706 and memory 708 arranged to configure the various elementsof the UE to perform the operations described herein.

In accordance with some embodiments, the MAC circuitry 704 may bearranged to contend for a wireless medium configure frames or packetsfor communicating over the wireless medium and the LTE physical layer(PHY) circuitry 702 may be arranged to transmit and receive signals. ThePHY 702 may include circuitry for modulation/demodulation,upconversion/downconversion, filtering, amplification, etc. In someembodiments, the processing circuitry 706 of the device 700 may includeone or more processors. In some embodiments, two or more antennas may becoupled to the physical layer circuitry arranged for sending andreceiving signals. The physical layer circuitry may include one or moreradios for communication in accordance with cellular (e.g., LTE) andWLAN (e.g., IEEE 802.11) techniques. The memory 708 may be storeinformation for configuring the processing circuitry 706 to performoperations for configuring and transmitting HEW frames and performingthe various operations described herein.

In some embodiments, the UE 700 may be part of a portable wirelesscommunication device, such as a personal digital assistant (PDA), alaptop or portable computer with wireless communication capability, aweb tablet, a wireless telephone, a smartphone, a wireless headset, apager, an instant messaging device, a digital camera, an access point, atelevision, a medical device (e.g., a heart rate monitor, a bloodpressure monitor, etc.), or other device that may receive and/ortransmit information wirelessly. In some embodiments, the UE 700 mayinclude one or more of a keyboard, a display, a non-volatile memoryport, multiple antennas, a graphics processor, an application processor,speakers, and other mobile device elements. The display may be an LCDscreen including a touch screen.

The one or more antennas 701 utilized by the UE 700 may comprise one ormore directional or omnidirectional antennas, including, for example,dipole antennas, monopole antennas, patch antennas, loop antennas,microstrip antennas or other types of antennas suitable for transmissionof RF signals. In some embodiments, instead of two or more antennas, asingle antenna with multiple apertures may be used. In theseembodiments, each aperture may be considered a separate antenna. In somemultiple-input multiple-output (MIMO) embodiments, the antennas may beeffectively separated to take advantage of spatial diversity and thedifferent channel characteristics that may result between each ofantennas and the antennas of a transmitting station. In some MIMOembodiments, the antennas may be separated by up to 1/10 of a wavelengthor more.

Although the UE 700 is illustrated as having several separate functionalelements, one or more of the functional elements may be combined and maybe implemented by combinations of software-configured elements, such asprocessing elements including digital signal processors (DSPs), and/orother hardware elements. For example, some elements may comprise one ormore microprocessors, DSPs, application specific integrated circuits(ASICs), radio-frequency integrated circuits (RFICs), radio-frequencyintegrated circuits (RFICs) and combinations of various hardware andlogic circuitry for performing at least the functions described herein.In some embodiments, the functional elements may refer to one or moreprocesses operating on one or more processing elements.

Embodiments may be implemented in one or a combination of hardware,firmware and software. Embodiments may also be implemented asinstructions stored on a computer-readable storage medium, which may beread and executed by at least one processor to perform the operationsdescribed herein. A computer-readable storage medium may include anynon-transitory mechanism for storing information in a form readable by amachine (e.g., a computer). For example, a computer-readable storagemedium may include read-only memory (ROM), random-access memory (RAM),magnetic disk storage media, optical storage media, flash-memorydevices, and other storage devices and media. In these embodiments, oneor more processors may be configured with the instructions to performthe operations described herein.

In some embodiments, the UE 700 may be configured to receive OFDMcommunication signals over a multicarrier communication channel inaccordance with an OFDMA communication technique. The OFDM signals maycomprise a plurality of orthogonal subcarriers. In some broadbandmulticarrier embodiments, eNBs may be part of a broadband wirelessaccess (BWA) network communication network, such as a WorldwideInteroperability for Microwave Access (WiMAX) communication network or a3rd Generation Partnership Project (3GPP) Universal Terrestrial RadioAccess Network (UTRAN) Long-Term-Evolution (LTE) or aLong-Term-Evolution (LTE) communication network, although the scope ofthe invention is not limited in this respect. In these broadbandmulticarrier embodiments, the UE 700 and the eNBs may be configured tocommunicate in accordance with an orthogonal frequency division multipleaccess (OFDMA) technique.

EXAMPLES AND ADDITIONAL NOTES

In Example 1, a method of calibrating a transmitter system parameterscan include receiving phase (φ) information derived from transmissioninformation in a transmitter, deriving an Inphase local oscillator(LO_I) signal and a quadrature phase local oscillator (LO_Q) signal froma combination of a first signal, which may be or include a tuningsignal, and the (φ) information, or a function of the φ information,receiving an RF signal provided by the transmitter, mixing the LO_Isignal and the LO_Q signal with the RF signal to obtain mixer outputsignals, and determining, as a function of the mixer output signals, theRF path delay and the IQ phase imbalance caused by the Polar TX.

In Example 2, the method of Example 1 optionally includes thetransmitter being a polar transmitter (Polar TX) including a phase lockloop (PLL), deriving at least AM information from at least one of themixer output signals, and concurrently determining the RF path delay andthe IQ phase imbalance by varying the first signal and measuringvariation of the derived AM information as the first signal is varied.

In Example 3, any one or more of Examples 1-2 optionally includesderiving at least AM information, and the variation of the AMinformation can comprise a substantially sinusoid function or anear-sinusoid function.

In Example 4, deriving AM information from at least one of the mixeroutput signals of any one or more of Examples 1-3 optionally includescoupling the mixer output signals to a CORDIC and obtaining the AMinformation as an output from the CORDIC.

In Example 5, determining the RF path delay of any one or more ofExamples 1-4 optionally includes calculating the difference in amplitudeof the substantially sinusoid function at two points of variation of thefirst signal and dividing the difference by a function of the firstsignal.

In Example 6, the first signal of any one or more of Examples 1-5 canoptionally comprise an FM signal (FM_tune), and determining the RF pathdelay can optionally include calculating:

$\text{RF Path Delay} = {{\Delta\; t} = \frac{\lbrack {( {2*\Pi*f_{R\; F}*\Delta\; t} ) - ( {( {2*\Pi*f_{R\; F}*\Delta\; t} ) + ( {2*\Pi*{FM\_ tune}*\Delta\; t} )} )} \rbrack}{( {2*\Pi*{FM\_ tune}} )}}$

where

-   -   f_(RF) is the frequency of the RF signal, and    -   FM_tune is the magnitude of FM_tune at each the two points.

In Example 7, the determining the IQ phase imbalance of any one or moreof Examples 1-6 optionally includes increasing the value of FM_tune infour discrete points for a full period of the AM information, measuringthe AM information as a function of each point, and calculating themagnitude |e| of the phase imbalance in terms of power as:

$\begin{matrix}{P_{aver} = {\frac{1}{4}{\sum\limits_{n = 1}^{4}A_{n}^{2}}}} \\{P_{{norm}\; 1} = {A_{n = 1}^{2} - P_{aver}}} \\{P_{{norm}\; 2} = {A_{n = 2}^{2} - P_{aver}}} \\{P_{\max} = \sqrt[2]{P_{{norm}\; 1}^{2} + P_{{norm}\; 2}^{2}}} \\{{{\mathbb{e}}} \approx \frac{P_{\max}}{P_{aver}}}\end{matrix}$

where A_(n) is the amplitude of the AM information at discrete point n.

In Example 8, the determining the IQ phase imbalance of any one or moreof Examples 1-7 can optionally include calculating the sign of the IQphase imbalance by comparing A_(n) measured at sequential quadrants of aconstellation diagram for the variation of the AM information.

In Example 9, the (LO_I) signal and the (LO_Q) signal of any one or moreof Examples 1-8 can optionally be derived using an analog phase log loopor a digital phase lock loop.

In Example 10, the RF signal that is provided by the Polar TX in any oneof Examples 1-9 can optionally be received by a feedback receiver (FBR).

In Example 11, polar transmitter (Polar TX) system can include a PolarTX comprising a radio frequency digital to analog converter (RFADC)configured to receive phase modulation information and processedamplitude information and to provide a radio frequency transmissionsignal for broadcast using an antenna and a phase lock loop (PLL)configured to receive both a function of the phase information and afirst signal, and to provide a function of the phase modulationinformation, an Inphase local oscillator (LO_I) signal and a quadraturephase local oscillator (LO_Q) signal, and a feedback receiver (FBR)comprising a plurality of mixers to mix the provided radio frequencytransmission signal with the (LO_I) signal and the (LO_Q) signal toobtain mixer output signals, and measuring circuitry configured toconcurrently determine, as a function of the mixer output signals, theRF path delay and the IQ phase imbalance caused by the Polar TX.

In Example 12, the Polar TX system of he measuring circuitry of Example11 can optionally derive at least AM information from at least one ofthe mixer output signals, and can optionally concurrently determine theRF path delay and the IQ phase imbalance by varying the first signal andmeasuring the variation of the AM information as the first signal isvaried.

In Example 13, the variation of the AM information of any one or more ofExamples 11-12 can optionally be a substantially sinusoid function orsinusoid-like function.

In Example 14 the deriving AM information from at least one of the mixeroutput signals of any one or more of Examples 11-13 can optionallyinclude coupling the mixer output signals to a coordinate rotationdigital computer (CORDIC) and obtaining the AM information as an outputfrom the CORDIC.

In Example 15 the determining the RF path delay of any one or more ofExamples 11-14 can optionally include calculating the difference inamplitude of the substantially sinusoid or sinusoid-like function at twopoints of variation of the first signal and dividing the difference by afunction of the first signal.

In Example 16 the first signal of any one or more of Examples 11-15 canoptionally be or include an FM signal (FM_tune), and determining the RFpath delay can optionally include calculating:

$\text{RF Path Delay} = {{\Delta\; t} = \frac{\lbrack {( {2*\Pi*f\; R\; F*\Delta\; t} ) - ( {( {2*\Pi*f\; R\; F*\Delta\; t} ) + ( {2*\Pi*{FM\_ tune}*\Delta\; t} )} )} \rbrack}{( {2*\Pi*{FM\_ tune}} )}}$

where

-   -   fRF is the frequency of the RF signal, and    -   FM_tune is the magnitude of FM_tune at each the two points.

In Example 17, determining the IQ phase imbalance of any one or more ofExamples 11-17 can optionally include increasing the value of FM_tune infour discrete points for a full period of the AM signal, measuring theAM information as a function of each point, and calculating themagnitude |e| of the phase imbalance in terms of power as:

$\begin{matrix}{P_{aver} = {\frac{1}{4}{\sum\limits_{n = 1}^{4}A_{n}^{2}}}} \\{P_{{norm}\; 1} = {A_{n = 1}^{2} - P_{aver}}} \\{P_{{norm}\; 2} = {A_{n = 2}^{2} - P_{aver}}} \\{P_{\max} = \sqrt[2]{P_{{norm}\; 1}^{2} + P_{{norm}\; 2}^{2}}} \\{{{\mathbb{e}}} \approx \frac{P_{\max}}{P_{aver}}}\end{matrix}$

where An is the amplitude of the AM information at point n.

In Example 18, the sign of the IQ phase imbalance of any one or more ofExamples 11-17 cam optionally be calculated by comparing An measured atsequential quadrants of a constellation diagram for the variation of theAM information.

In Example 19, the comparing of any one or more of Examples 11-18 canoptional be or include a subtraction process.

In Example 20, wherein the RF signal provided by the Polar TX of any oneor more of Examples 11-19 can optionally be received by the (FBR).

Example 21 can include, or can optionally be combined with any portionor combination of any portions of any one or more of Examples 1 through20 to include, subject matter that can include means for performing anyone or more of the functions of Examples 1 through 20, or amachine-readable medium including instructions that, when performed by amachine, cause the machine to perform any one or more of the functionsof Examples 1 through 20.

The above detailed description includes references to the accompanyingdrawings, which form a part of the detailed description. The drawingsshow, by way of illustration, specific embodiments in which theinvention can be practiced. These embodiments are also referred toherein as “examples.” All publications, patents, and patent documentsreferred to in this document are incorporated by reference herein intheir entirety, as though individually incorporated by reference. In theevent of inconsistent usages between this document and those documentsso incorporated by reference, the usage in the incorporated reference(s)should be considered supplementary to that of this document; forirreconcilable inconsistencies, the usage in this document controls.

In this document, the terms “a” or “an” are used, as is common in patentdocuments, to include one or more than one, independent of any otherinstances or usages of “at least one” or “one or more.” In thisdocument, the term “or” is used to refer to a nonexclusive or, such that“A or B” includes “A but not B,” “B but not A,” and “A and B,” unlessotherwise indicated. In the appended claims, the terms “including” and“in which” are used as the plain-English equivalents of the respectiveterms “comprising” and “wherein.” Also, in the following claims, theterms “including” and “comprising” are open-ended, that is, a system,device, article, or process that includes elements in addition to thoselisted after such a term in a claim are still deemed to fall within thescope of that claim. Moreover, in the following claims, the terms“first,” “second,” and “third,” etc. are used merely as labels, and arenot intended to impose numerical requirements on their objects.

The above description is intended to be illustrative, and notrestrictive. For example, the above-described examples (or one or moreaspects thereof) may be used in combination with each other. Otherembodiments can be used, such as by one of ordinary skill in the artupon reviewing the above description. Also, in the above DetailedDescription, various features may be grouped together to streamline thedisclosure. This should not be interpreted as intending that anunclaimed disclosed feature is essential to any claim. Rather, inventivesubject matter may lie in less than all features of a particulardisclosed embodiment. Thus, the following claims are hereby incorporatedinto the Detailed Description, with each claim standing on its own as aseparate embodiment. The scope of the invention should be determinedwith reference to the appended claims, along with the full scope ofequivalents to which such claims are entitled.

What is claimed is:
 1. A polar transmitter system comprising: a polartransmitter comprising a radio frequency digital to analog converter(RFDAC) to receive phase modulation information and processed firstamplitude modulation (AM) information to provide a radio frequencytransmission signal for broadcast, and oscillator circuitry configuredto receive both a function of the phase modulation information and of afirst signal, and to provide at least an Inphase local oscillator (LO_I)signal and a quadrature phase local oscillator (LO_Q) signal; and afeedback receiver (FBR) comprising a plurality of mixers to mix theradio frequency transmission signal with the LO_I signal and the LO_Qsignal to obtain a plurality of mixer output signals, processingcircuitry to derive an AM component and a phase component from theplurality of mixer output signals, and measuring circuitry to monitor atleast one of the AM component or the phase component, as the firstsignal is varied, to concurrently determine, both an RF path delay andan IQ phase imbalance, each as a function of both the monitored at leastone component and of the varied first signal.
 2. The polar transmitterof claim 1 wherein the oscillator circuitry comprises a phase lock loop(PLL).
 3. The polar transmitter (Polar TX) system of claim 2 wherein themeasuring circuitry further derives at least a second AM informationfrom at least one of the mixer output signals, varies the first signaland measures the variation of the second AM information as the firstsignal is varied to determine the RF path delay and the IQ phaseimbalance.
 4. The Polar TX system of claim 3 wherein the varied secondAM information comprises a substantially sinusoid trajectory.
 5. ThePolar TX system of claim 4 wherein the second AM information is derivedby coupling the mixer output signals to a coordinate rotation digitalcomputer (CORDIC), the second AM information obtained as an output fromthe CORDIC.
 6. The Polar TX system of claim 4 wherein the RF path delayis derived by calculating the difference in amplitude of thesubstantially sinusoid trajectory at two points of variation of thefirst signal and dividing the difference by a function of the firstsignal.
 7. The Polar TX system of claim 4 wherein the first signal is anFM signal (FM_tune), the RF path delay is Δt, and the RF path delay isdetermined by calculating:$\text{RF Path Delay} = {{\Delta\; t} = \frac{\lbrack {( {2*\Pi*f\; R\; F*\Delta\; t} ) - ( {( {2*\Pi*f\; R\; F*\Delta\; t} ) + ( {2*\Pi*{FM\_ tune}*\Delta\; t} )} )} \rbrack}{( {2*\Pi*{FM\_ tune}} )}}$where fRF is the frequency of the RF signal; and FM_tune is themagnitude of FM_tune at each of two points of the variation of the firstsignal.
 8. The Polar TX system of claim 4 wherein the first signal is anFM signal (FM tune) and the IQ phase imbalance is determined byincreasing the value of FM tune in a plurality of discrete points for afull period of variation of the second AM information, the second AMinformation is measured as a function of each of the plurality ofpoints, and the magnitude |e| of the IQ phase imbalance is calculated interms of power as: $\begin{matrix}{P_{aver} = {\frac{1}{4}{\sum\limits_{n = 1}^{4}A_{n}^{2}}}} \\{P_{{norm}\; 1} = {A_{n = 1}^{2} - P_{aver}}} \\{P_{{norm}\; 2} = {A_{n = 2}^{2} - P_{aver}}} \\{P_{\max} = \sqrt[2]{P_{{norm}\; 1}^{2} + P_{{norm}\; 2}^{2}}} \\{{{\mathbb{e}}} \approx \frac{P_{\max}}{P_{aver}}}\end{matrix}$ where An is the amplitude of the second AM information atpoint n.
 9. The Polar TX system of claim 8 wherein the sign of the IQphase imbalance is calculated by comparing An measured at sequentialquadrants of a constellation diagram for the variation of the second AMinformation.
 10. The Polar TX system of claim 9 wherein the comparing isa subtraction process.
 11. The Polar transmitter system of claim 2wherein the RF signal provided by the Polar TX is received by the FBR.12. A method of calibrating parameters in a polar transmitter (Polar TX)system comprising: receiving phase information derived from transmissioninformation in a Polar TX; deriving an Inphase local oscillator (LO_I)signal and a quadrature phase local oscillator (LO_Q) signal from acombination of a first signal and the phase information; receiving an RFsignal provided by the Polar TX; mixing the LO_I signal and the LO_Qsignal with the RF signal to obtain mixer output signals; derivingamplitude modulation (AM) information and phase information from themixer output signals; and concurrently determining, by monitoring atleast the amplitude of the derived AM information, as the first signalis varied, both an RF path delay and an IQ phase imbalance in the PolarTX system, each as a function of the monitored derive AM information andof the varied first signal.
 13. The method of claim 12 wherein thevarying the first signal results in the amplitude of the derived AMinformation having a substantially sinusoid trajectory.
 14. The methodof claim 13, wherein determining the RF path delay comprises calculatingthe difference in amplitude of the derived AM information having thesubstantially sinusoid trajectory at two points of variation of thefirst signal and dividing the difference by a function of the firstsignal.
 15. The method of claim 14 wherein the first signal is an FMsignal (FM_tune), the RF path delay is Δt, and determining the RF pathdelay comprises calculating:${{RF}\mspace{14mu}{Path}\mspace{14mu}{Delay}} = {{\Delta\; t} = \frac{\lbrack {( {2*\Pi*{fRF}*\Delta\; t} ) - ( {( {2*\Pi*{fRF}*\Delta\; t} ) + ( {2*\Pi*{FM\_ tune}*\Delta\; t} )} )} \rbrack}{( {2*\Pi*{FM\_ tune}} )}}$where fRF is the frequency of the RF signal; and FM_tune is themagnitude of FM_tune at each of the two points.
 16. The method of claim13 wherein the first signal is an FM signal (FM_tune) and determiningthe IQ phase imbalance comprises increasing the value of FM_tune in aplurality of discrete points for a full period of the derived AMInformation, measuring the derived AM information as a function of eachof the plurality of discrete points, and calculating the magnitude |e|of the IQ phase imbalance in terms of power as: $\begin{matrix}{P_{aver} = {\frac{1}{4}{\sum\limits_{n = 1}^{4}A_{n}^{2}}}} \\{P_{{norm}\; 1} = {A_{n = 1}^{2} - P_{aver}}} \\{P_{{norm}\; 2} = {A_{n = 2}^{2} - P_{aver}}} \\{P_{\max} = \sqrt[2]{P_{{norm}\; 1}^{2} + P_{{norm}\; 2}^{2}}} \\{{{\mathbb{e}}} \approx \frac{P_{\max}}{P_{aver}}}\end{matrix}$ where An is the amplitude of the AM information atdiscrete point n.
 17. A cellular transceiver comprising: a polartransmitter (Polar TX) to receive phase modulation information andprocessed first amplitude modulation (AM) information to provide a radiofrequency transmission signal for broadcast, the Polar TX comprising aphase lock loop (PLL) to receive both a function of the phase modulationinformation and of a first signal, to provide at least an Inphase localoscillator (LO_I) signal and a quadrature phase local oscillator (LO_Q)signal; and a feedback receiver (FBR) comprising a plurality of mixersto mix the radio frequency transmission signal with the LO_I signal andthe LO_Q signal to obtain mixer output signals, processing circuitry toderive an AM component and a phase component from the plurality of mixeroutput signals, and measuring circuitry configured to monitor at leastone of the AM component or the phase component, as the first signal isvaried, to concurrently determine an RF path delay and an IQ phaseimbalance, each as a function of both the monitored at least onecomponent and of the varied first signal.
 18. The cellular transceiverof claim 17 wherein the measuring circuitry further derives at leastsecond AM information from at least one of the mixer output signals, andconcurrently varies the first signal and measures the variation of thesecond AM information as the first signal is varied to determine the RFpath delay and the IQ phase imbalance.
 19. The cellular transceiver ofclaim 18 wherein the variation of the second AM information comprises asubstantially sinusoid trajectory.
 20. The cellular transceiver of claim19 wherein the second AM information is derived by coupling the mixeroutput signals to a coordinate rotation digital computer (CORDIC) andobtaining the second AM information as an output from the CORDIC. 21.The cellular transceiver of claim 19 wherein the RF path delay isderived by calculating the difference in amplitude of the substantiallysinusoid trajectory at two points of variation of the first signal anddividing the difference by a function of the first signal.
 22. Thecellular transceiver of claim 21 wherein the first signal is an FMsignal (FM_tune), the RF path delay is Δt, and the RF path delay isdetermined by calculating:$\text{RF Path Delay} = {{\Delta\; t} = \frac{\lbrack {( {2*\Pi*f\; R\; F*\Delta\; t} ) - ( {( {2*\Pi*f\; R\; F*\Delta\; t} ) + ( {2*\Pi*{FM\_ tune}*\Delta\; t} )} )} \rbrack}{( {2*\Pi*{FM\_ tune}} )}}$where fRF is the frequency of the RF signal; and FM_tune is themagnitude of FM_tune at each of the two points.
 23. The cellulartransceiver of claim 19 wherein the first signal is an FM signal(FM_tune) and the IQ phase imbalance is determined by increasing thevalue of FM_tune in a plurality of discrete points for a full period ofvariation of the second AM information, measuring the second AMinformation as a function of each of the plurality of points, andcalculating the magnitude |e| of the IQ phase imbalance in terms ofpower as: $\begin{matrix}{P_{aver} = {\frac{1}{4}{\sum\limits_{n = 1}^{4}A_{n}^{2}}}} \\{P_{{norm}\; 1} = {A_{n = 1}^{2} - P_{aver}}} \\{P_{{norm}\; 2} = {A_{n = 2}^{2} - P_{aver}}} \\{P_{\max} = \sqrt[2]{P_{{norm}\; 1}^{2} + P_{{norm}\; 2}^{2}}} \\{{{\mathbb{e}}} \approx \frac{P_{\max}}{P_{aver}}}\end{matrix}$ where An is the amplitude of the second AM information atpoint n.
 24. The cellular transceiver of claim 23 wherein the sign ofthe IQ phase imbalance is calculated by comparing An measured atsequential quadrants of a constellation diagram for the variation of thesecond AM information.